Direct digitally tunable microwave oscillators and filters

ABSTRACT

A tunable element in the microwave frequency range is described that may include one or more tunable elements that are directly digitally controlled by a digital bus connecting a digital control circuit to each controlled element. In particular, each digital signal is filtered by a digital isolation technique so that the signal reaches the tunable elements with very low noise. The low noise digital signals are then converted to analog control voltages. The direct D/A conversion is accomplished by a special D/A converter which is manufactured as an integral part of a substrate. This D/A converter in accordance with the invention may consist of a resistor ladder or a directly digitally controlled capacitor. The direct digitally controlled capacitor may be a cantilevered type capacitor having multiple separate electrodes or sub-plates representing binary bits that may be used to control the capacitor. A low cost microwave oscillator is disclosed in which some of the filters and oscillators are direct digitally tuned elements.

This is a divisional of application Ser. No. 09/376,867, filed on Aug.18, 1999 now U.S. Pat. No. 6,741,449, entitled DIRECT DIGITALLY TUNABLEMICROWAVE OSCILLATORS AND FILTERS.

BACKGROUND OF THE INVENTION

This invention relates generally to a communications device and inparticular to direct digitally tunable microwave oscillators and filtersthat enable a low-cost high-speed digital transceiver in the millimeterwave frequency range.

Telecommunication deregulation and the growth of the Internet arecausing a large demand for broadband communications network access tohomes and offices at increasing bit rates. For Internet access, aDigital Subscriber Loop (DSL) service and similar techniques arebecoming popular. However, the speed of the DSL service is limited bythe fact that the data travels over long telephone twisted copper wirepairs. Cable modems are also becoming prevalent methods of Internet andtelephony access. However, the return channel for a cable modem is quitelimited in speed by the nature of cable distribution technology. Fiberoptics offers higher access speeds than the other systems and services,but fiber does not currently reach most homes and offices andinstallation of new fiber is very expensive.

To overcome these limitations with current systems, service providersand users are considering using digital wireless communications as apossible alternative access system. Various radio frequency bands are infact currently allocated to applications that include broadband wirelessaccess, however, most of the available bandwidth is in the millimeterwave range, ranging roughly from 15 to 66 GHz. Among these bands, theUSA Local Multipoint Distributed Services (LMDS) bands in the 28 to 30GHz range are a good example of a licensed band intended for telecomaccess applications. Since LMDS is a licensed band, the license ownerhas the exclusive rights to use the band for such services, thusensuring interference-free operation.

LMDS services may use point to multipoint communication networks. TheLMDS service provider typically maintains base station antennas on tallstructures to maintain line of sight to a large number of user-buildingsin a sector. Each subscribed user gets a small transceiver installedoutdoors. While the base station cost is divided among many users, thesubscriber transceiver serves only one user or at the most few users ina shared building. Thus, the cost of the transceiver must be kept lowfor the LMDS service to be economical. The option of reducing the costof the transceiver by using a lower frequency is not always practicalsince it may be precluded by spectrum availability limitations so thatmillimeter wave transceivers are desirable, but too expensive for manyapplications. To make millimeter wave communications cost effective, adrastic cost reduction of the millimeter wave transceivers is required.

A typical millimeter wave transceiver includes a frequency synthesizerthat generates the final millimeter wave frequency with an offset of afew GHz depending on the particular radio application. The synthesizeris required for accurately setting the exact transmit and receivefrequencies of the communications. An undesirable by-product offrequency synthesis is phase noise that must be kept within acceptableperformance levels for a particular radio link. Furthermore, asynthesizer may require a large tuning range, usually a few hundred MHz,to allow it to tune to alternate channels. A common way of implementinga synthesizer is to use a base phase locked loop at a lower frequency,such as around 2 GHz, and a chain of frequency multipliers. For example,a 28 GHz signal may be synthesized from a 1.75 GHz signal multiplied by16 (i.e., 2×2×2×2) which may be implemented using a chain of fourfrequency doublers.

There are several limitations to the above typical frequencysynthesizer. The cost of the synthesizer is significant because thefrequency doublers add complexity and cost. Furthermore, the synthesizeris very sensitive to noise. In particular, the phase locked-loop in thesynthesizer has a voltage controlled oscillator (VCO). The tuningvoltage of that VCO causes the final frequency of the VCO to vary by afew MHz so that one millivolt of noise in the phase locked loop controlvoltage may cause a frequency deviation of 500 kHz. This frequencydeviation may cause temporary loss of frequency lock and large biterrors in the communications data stream which are both undesirable.

Apart from the frequency synthesizer, the millimeter-wave filters in atransceiver also increase the cost of the transceiver, especially infrequency division duplex (FDD) transceivers in which a diplexer isused. The above is also true for a time division duplex (TDD) receiverwhich also needs filters. Thus, it is desirable to provide a directdigitally tunable oscillator and filter that overcomes the abovelimitations and problems with typical oscillators and filters that maybe used in millimeter wave transceiver devices and systems and it is tothis end that the present invention is directed.

SUMMARY OF THE INVENTION

The tunable filters and oscillators in the microwave frequency range inaccordance with the invention may include one or more tunable elementsthat are directly digitally controlled by a digital bus connecting adigital control circuit to each controlled element. In particular, eachdigital signal is filtered by a digital isolation technique so that thesignal reaches the tunable elements with very low noise. The low noisedigital signals are then converted to analog control voltages. Thedirect D/A conversion is accomplished by a special D/A converter whichis manufactured as an integral part of a substrate. This D/A converterin accordance with the invention may consist of a resistor ladder or adirectly digitally controlled capacitor.

The digitally controlled capacitor is tuned by electrostatic attractionin which one of the capacitor plates may bend towards a control plate ora set of control plates that causes an increase in the capacitance ofthe capacitor. The digital control in accordance with the invention maybe achieved by partitioning the set of control plates into a pluralityof sub-plates whose positions and dimensions (area) affect the overallcapacitance of the capacitor such that each sub-plate represents asingle weight of a weighted binary sum. In other words, a larger platemay have an area A (equal to one half the size of the other plate of thecapacitor), while a smaller plate has an area of A/2, a still smallerplate has an area of A/4 and a smallest plate has an area of A/8.

To control/tune the capacitor to a desired capacitance, one or more ofthe plates are charged with low noise voltages (Vcc or 0) to produce thedesired capacitance. For example, if only the largest plate is charged,the capacitance of the capacitor may be about ½ of its total value. Itthe first and second plates are energized (A+A/2), then the capacitorhas a capacitance equal to about ¾ of its total value. In this manner,each plate represents a binary weight (1, ½. ¼, ⅛, etc.) that may beused to control the capacitor. Any final fine-tuning of the capacitormay involve a small sub-plate driven by an analog voltage. Thecontrolling of the capacitor in turn is used to tune a microwave circuitsuch as a oscillator or filter.

Various different capacitor structures are possible in accordance withthe invention. In accordance with the invention, these capacitorstructures are modified micro-machined parallel plate cantilevercapacitors or interdigital capacitors. In addition, various combinationsand orientations of the set of sub-plates relative to the other plateare possible to accomplish the desired weight distribution of thedigital control word in accordance with the invention. For example, theset of sub-plates may be positioned perpendicular to the cantileverplate or parallel to the cantilever plate in cantilever capacitor inaccordance with the invention.

These digitally controlled capacitors in accordance with the inventionmay be used to control the frequency of an oscillator or of a filter,especially in the microwave frequency range. By combining a digitaltuner, a resonator and active microwave devices, a digitally controlledoscillator (DCO) is accomplished. The DCO in accordance with theinvention with proper digital noise isolation may be connected to adigital circuit simplifying the implementation of a microwave frequencysynthesizer by eliminating the analog control loop, yet allowing widetuning range with minimum noise.

The digitally controlled devices in accordance with the invention mayalso be used to implement other electrically tuned devices, such assemiconductor varactor diodes. The low noise digital tuning may beaccomplished by implementing a resistor ladder type of D/A converter inaccordance with the invention that requires no active devices betweenthe noise-reduced digital control word and the controlled device. Theresistors may be deposited on a metal layer that acts as a shield fromthe circuit substrate. This can be accomplished by various well knownthin film techniques. To further reduce noise, some of these resistorswhich are connected to the control bits may be placed outside a shieldedmicrowave circuit containing the tunable element, and are fed to theshielded region via bypass capacitors, forming and RC low pass filter.

With direct digitally tuned microwave elements in accordance with theinvention, a low cost microwave transceiver is disclosed in which someof the filters and oscillators are direct digitally tuned to reducenoise. The tuning is possible even in the transceiver front-enddiplexer. The tuning in accordance with the invention may reduce thenumber of required diplexer filtering stages and may be used also totune away undesired signals that are not rejected by a conventional bandpass filter inside a diplexer.

In accordance with another aspect of the invention, the partitioning ofthe sub-plates by size (area) and position may be done with a method inaccordance with the invention that includes the steps of placing a firstsub-plate of a desired weight-effect where desired and adjusting thewidth of this plate until the desired weight-effect associated with thissub-plate is achieved and confirmed by measurement or simulation. Then asecond sub-plate is placed in the presence of the first plate and itsdimensions are adjusted until the desired weight-effect associated withthis second sub-plate is achieved and confirmed by measurement orsimulation. This process is repeated for all remaining plates to achievethe direct digitally tuned element in accordance with the invention.

Thus, in accordance with the invention, a direct digitally controlledmicrowave tuning element is provided comprising a microwave circuit thatis being tuned to a predetermined microwave frequency and anelectrically tunable element for tuning the microwave circuit to thepredetermined microwave frequency. The electrically tunable element maybe attached to a substrate and may further comprise a plurality oftuning signals, means for noise filtering said tuning signals togenerate noise filtered tuning signals, a converter, based on the noisefiltered tuning signals, that controls the frequency of the microwavecircuit to a predetermined microwave frequency and an analog tuningsignal that is integrated into the substrate and mounted on said noisefiltering means.

In accordance with another aspect of the invention, a direct digitallycontrolled element for tuning to a microwave frequency is providedwherein the element comprises means for generating one or more digitalsignals, means for noise isolating the one or more digital signals inorder to reduce the noise contained in the one or more digital signals,and means for controlling a device using the noise isolated digitalbinary signals, the device changing a predetermined characteristic inresponse to the digital signals so that the device tunes itself to amicrowave frequency based on the digital signals.

In accordance with yet another aspect of the invention, a directdigitally controlled capacitor for tuning a circuit to a microwavefrequency is provided wherein the capacitor comprises a control platethat deflects in response to a second plate being charged and aplurality of tuning signals attached to the sub-plates of the capacitorfor controlling the capacitor. The second plate further comprises one ormore sub-plates electrically isolated at DC or low frequencies from eachother and from said moving plate, the sub-plates controlling thedeflection of the moving plate in order to change the microwavefrequency response of the capacitor. In accordance with yet anotheraspect of the invention, a digital microwave transceiver is providedcomprising means for receiving a signal to be transmitted using amicrowave frequency, and means for modulating the signal onto apredetermined microwave frequency, the modulator comprising one or moredirect digitally tuned circuits for precisely controlling thepredetermined microwave frequency of the modulator.

In accordance with yet another aspect of the invention, a method fordetermining the sub-plates areas of one or more sub-plates in a tuningdevice is provided, comprising positioning a first sub-plate of alargest weight-effect in a predetermined location, adjusting thedimensions of said first sub-plate until the desired weight-effectassociated with said sub-plate is achieved, positioning a secondsub-plate of a smaller weight-effect adjacent the first sub-plate, andadjusting the dimensions of the second sub-plate until the desiredweight-effect associated with said second sub-plate is achieved.

In accordance with other aspects of the invention, a direct digitallycontrolled oscillator is provided wherein the oscillator comprises atunable oscillator circuit, and a direct digitally tuned circuitconnected to the oscillator circuit for controlling the frequency of theoscillator circuit. A direct digitally tunable filter is also providedwherein the filter comprises at least one resonator element, and atleast one direct digitally tuned circuit electrically coupled to saidresonator to control the frequency of the resonator.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a block diagram of a typical radio access terminal including aradio transceiver module portion;

FIG. 2 is a block diagram of a typical radio access terminal dividedinto indoor and outdoor units with a digital interface between them;

FIG. 3 is a block diagram illustrating an example of a typical radiotransceiver module;

FIG. 4 illustrates a typical digital phase locked loop;

FIG. 5 illustrates a digital phase locked loop with noise isolation ofthe digital signals and a digitally controlled oscillator in accordancewith the invention;

FIG. 6 illustrates an example of a digitally controlled oscillator inaccordance with the invention with noise isolation by optical coupling;

FIG. 7 a illustrates an example of a digitally controlled oscillator inaccordance with the invention with noise isolation using a resistor anda feed-through capacitor;

FIG. 7 b illustrates an example of a noise-isolated direct digitallycontrolled microwave circuit in accordance with the invention;

FIG. 8 illustrating an example of a digitally controlled capacitor inaccordance with the invention with a moving cantilever beam andsub-plates arranged in parallel to the beam;

FIG. 9 illustrates a digitally controlled capacitor in accordance withthe invention with a moving cantilever beam and sub-plates arrangedperpendicularly to the beam;

FIG. 10 illustrates a plate arrangement of a digitally controlledcapacitor in accordance with the invention with a moving plate based onan interdigital cantilever beam;

FIG. 11 illustrates an example of a plate arrangement of a digitallycontrolled capacitor in accordance with the invention with a movingplate based on an interdigital cantilever beam and sub-plates arrangedperpendicularly to the moving interdigital beam;

FIG. 12 illustrates a cross section of a cantilever beam with digitalcontrol sub-plates mounted on a substrate with other components inaccordance with the invention;

FIG. 13 illustrates a digitally tuned capacitor in accordance with theinvention electrically connected to a microwave resonator;

FIG. 14 illustrates a noise-isolated direct digitally controlledmicrowave circuit in accordance with the invention with aresistor-ladder D/A converter;

FIG. 15 illustrates a cross section through a shielding structure of thenoise-isolated direct digitally controlled microwave circuit of FIG. 14;and

FIG. 16 illustrates a block diagram of a digital microwave transceiverincluding direct digitally tuned circuits in accordance with theinvention.

DETAILED DESCRIPTION OF A PREFERRED EMBODIMENT

The invention is particularly applicable to digitally controlledoscillators and filters for use in a microwave transceiver or microwaveaccess terminal and it is in this context that the invention will bedescribed. It will be appreciated, however, that the device and methodin accordance with the invention has greater utility, such as to othercommunications systems that require a tuned oscillator or filter. Beforedescribing the invention, a typical microwave subscriber terminal thatmay include a digitally controlled filter or oscillator in accordancewith the invention will be described.

A typical digital microwave transceiver 100 for a subscriber terminal ina fixed wireless network is shown as a block diagram in FIG. 1. Inparticular, FIG. 1 shows an integral outdoor transceiver 100 while FIG.2 shows a split transceiver 200, including an indoor unit 210, anoutdoor unit 220 and a connecting cable 201 between them. The integralunit of FIG. 1 includes a user interface 101 for interfacing between thecommunications system and the microwave transmission and receptionsystem and converting user traffic to a serial bitstream fortransmission over the air, an air-interface media access control layer(MAC) 102 if applicable for the wireless network in use, a systemcontroller 103, for configuration, protocol processing and networkmanagement tasks and a antenna 110 for receiving/transmitting themicrowave signals. The transceiver-related functions more directlyrelated to the present invention are depicted inside a block 111 havinga gray background. These functions are included in a modem 104 forconverting the serial bit stream into modulated signals and vice versa,an up/down frequency converter 105 for converting between differentfrequencies, a synthesizer 106 for setting the proper radio frequencyfor communication and a millimeter wave front-end 107 that performs thehigher frequency transceiver functions. The control signals betweenthese units and the system controller are shown although not describedsince these are well known. The functions depicted in FIG. 1 are wellknown and typical for a microwave transceiver and will not be describedhere in any more detail. The separation of the transmit-receivefunctions, the number of conversion stages, the DC power supply andother radio-related features are omitted from the figure since these arealso well known in the art.

FIG. 2 illustrates a split transceiver 200 including an indoor unit 210and an outdoor unit 220. The split transceiver of FIG. 2 includes thesame blocks as the transceiver shown in FIG. 1 and further includes anindoor to outdoor connection medium 201 that may be a coaxial cable, afiber optic cable, a twisted pair cable or combinations of such media.An integral transceiver is likely to be more cost-effective, howevervarious constraints in the field, such as distance from the user to theoutdoor unit, may necessitate a split implementation. In a splittransceiver, one typical partition between the indoor unit and theoutdoor unit is to keep the modem indoors and use intermediate frequency(IF) signals for indoor to outdoor transmission. A lower cost typicalapproach is shown in FIG. 2 in which the modem 104 is outdoors and themedia 201 transmits multiplexed digital information to the modem using adigital multiplexer 203 and a media physical layer driver (Media PHY)204. The multiplexed signal contains the data, timing reference andcontrol information. For example, a transmission may consist of astart-stop protocol as used in a serial port of a UART. The start bitindicates when a transmission in the air should start. A media PHYdriver 207 and multiplexer 205 in the outdoor unit 220 can interpret thestop bit and control the modem 104 via a transmit control bus 206. Themedia PHY driver can be an off the shelf transceiver for the appropriatemedium, such as 100BaseT driver for twisted pair cabling. Now, moredetails of the portion of the transceiver in the gray background will bedescribed.

FIG. 3 illustrates the portion 111 of the transceiver in more detail. Inparticular, data for transmission enters a Digital Processor 301 whichincludes a modem. In order to transmit a signal, the well known I and Qsamples of the modulation are output via two D/A converters 302, 303followed by smoothing filters (not shown) as required which aremodulating a sine wave generated by a phase locked loop (PLL) 304 atabout 1 GHz. A second conversion occurs using a voltage-controlledoscillator (VCO) 305 and a mixer 306. The VCO 305 is operating at thefinal local oscillator frequency, such as 28 GHz. The output frequencyof the mixer 306 and bandpass filter 307 is thus 29 GHz, if the filter307 is tuned to reject the other image at 27 GHz. A power amplifier 320drives the signal via a diplexer 308 to the antenna port 309.

To receive incoming signals, a receiving chain includes the other branchof the diplexer 308, a low noise amplifier 310, a bandpass filter 311and a mixer 312. The receiving frequency may be 30 GHz, assumingtransmit-receive frequency difference of 1 GHz. Therefore the receivingIF frequency is 2 GHz, and a receiving PLL 314 runs at this frequency. Aquadrature demodulator 315 converts the frequency to baseband and thesignals are digitized by an A/D converter 313, including anyanti-aliasing filters (not shown) as required.

A transceiver also includes AGC functions and other functions, which areknown in the art but are not relevant to this disclosure. The VCO 305 isa part of a synthesizer that tunes the transceiver. The VCO in thistransceiver has noise problems which will be described below that keepthe cost of the transceiver high since expensive VCO are needed tomaintain the necessary spectral purity. The VCO forms a PLL with aprescaler 316, a digital circuit for phase detector and loop filterwithin the digital processor 301 and a D/A converter 317, also shown inFIG. 4 and described below. When receiving signals, the digitalprocessor 301 demodulates the received signals and delivers receiveddata to the other functions shown in FIGS. 1 and 2. The synthesizer inthe digital processor is supposed to be locked to a frequency reference318, which may be a crystal oscillator.

In some applications, to further save cost, the prescaler 316 is omittedand the synthesizer is phase locked to the received signal via the modemcarrier recovery loop in the digital processor 301. Other optionsinclude running the VCO at 1/n of the desired frequency, where n is asmall integer such as 2 and 3, and the mixers 306, 312 are harmonicmixers. The above typical transceiver architecture is based on frequencydivision duplex (FDD) as is well known. A time division duplex variantis also possible and well known, wherein the diplexer 308 is replaced bya transmit/receive switch and a single bandpass filter, and making thePLLs 304, 314 the same frequency.

FIG. 4 illustrates a noise problem inherent in a typical VCO 400 thatlimits the spectral purity of the VCO and thus limits the performance ofa microwave transceiver that uses the VCO. As shown in FIG. 4, afeedback loop may be formed with the VCO to control the frequency of theVCO. The feedback loop may include a digital phase detector 402, adigital loop filter 403, a D/A converter 404 and a divide by N (/N)circuit 405. In operation, the signals output from the VCO aredownconverted by the /N converter and fed into the digital phasedetector. The digital phase detector compares the frequency of the VCOto a standard and generates a control signal to adjust the VCO. Thecontrol signal is passed through the digital loop filter to reduce noiseand then converted into an analog control signal by the D/A converter.The analog control signal is fed into the VCO to adjust the signaloutput from the VCO.

In more detail, a voltage control line 401 frequency-modulates the VCO.Thus, if the VCO is tunable across 500 MHz, and the voltage controlrange is between 0V and 5V, the VCO's sensitivity is 100 MHz/volt.Therefore, a noise spike of one millivolt (very small) will cause aabrupt 500 kHz frequency deviation that may cause frequency lock to belost in a microwave transceiver. This noise may come from ground loopsor noise induced on the line 401. To overcome this noise problem, adirect digital controlled VCO in accordance with the invention isintroduced. The direct digitally controlled VCO in accordance with theinvention will now be described.

FIG. 5 illustrates a direct digitally controlled oscillator 500 inaccordance with the invention that may also be referred to as adigitally-controlled oscillator (DCO) 501. The digital control word 505for the DCO may be noise-isolated by a noise isolation circuit 502, andthe noise-free digital word 505 may drive the DCO to produce aparticular frequency signal while reducing undesirable noise. Inaccordance with the invention, it is also possible to split the digitalcontrol word 505 so that the coarse tuning may be completed andcontrolled by a set of most significant bits 503, and the fine tuning ofthe DCO may be completed and controlled using a signal 504 that may beeither digital or analog. The analog fine tuning will require a D/Aconverter and it has the same noise level as the original VCO controlline 401, but the DCO sensitivity of this signal is much lower. Forexample, if a total of 8 bits is used, and the least significant bit iscontrolling the DCO as an analog line ranging between 0V and 5V, thesensitivity of this line is 1/256 of the combined original thus thenoise is attenuated by 256, or 48 dB. If the digital signals are heldfixed for a receiving session and the analog line is the only one tuned,the lower loop gain in the PLL further simplified the loop design andreduced the noise sensitivity of the DCO.

The number of digital bits that control the DCO is set so that thecoarse setting covers all instabilities and noise except the temperatureinstability range of the oscillator 501 and the analog range covers thatinstability. If the frequency lock of the DCO 501 is lost, the digitalprocessor 301 (shown in FIG. 3) initiates a coarse search by varying thedigital signal 503 to achieve a coarse frequency lock and then uses thefine tuning signal 504 for final lock. If the number of availabledigital bits is large enough (to minimize the quantization noise betweeneach digital bit), an all-digital controlled loop is possible. Inparticular, the lease significant digital bit defines the increment offrequency error (range of frequencies) and the DCO will wander betweenthese frequencies. Furthermore, in an all-digital implementation, thetwo signals 503 and 504 may have some overlapping range. For example,the coarse tuning signal 503 may set the frequency at steps of 20 MHz,but the fine tuning signal 504 range is up to 50 MHz. Such overlap ispossible because the weights of each fine tuning bit is not necessarilyanother power of 2. For example, the digital bits of the coarse signal503 may have weights of 128, 64, 32, 16 and 8 and the fine bus may haveweights 14,8,4,2,1 wherein the weight of 14 is an example of non-powerof 2. In operation, the DCO frequency, assuming linear controlcharacteristics, is therefore proportional to the sum of the weights forall the digital bits set to 1. For example, with the above order ofbits, the control words on the signals 503, 504 may have a value of“1000010001” may set a frequency of the DCO 501 proportional to128+14=142 which means that the actual frequency of the DCO will be alittle above the halfway point (142/255) of the tuning range. If the DCOcontrol is not linear, the frequency is still a monotonic function ofthe sum of weights if the DCO is well designed. Now, the noise isolationin accordance with the invention will be described in more detail.

FIGS. 6 and 7 a illustrate two methods in accordance with the inventionfor performing noise isolation. In FIG. 6, an all digital signals may beopto-coupled to the DCO 603 through a shielded enclosure 604, and thedigital levels are drawn from an analog supply voltage VCC 601 and alocal analog ground 602. In FIG. 7, one or more isolation resistors 701may provide the desired noise isolation by performing low-pass RCfiltering with a feed-through capacitance 702 of a shield 703surrounding the DCO 704. Now, a direct digitally tuned microwave elementin accordance with the invention will be described in more detail.

FIG. 7 b illustrates a direct-digitally tuned microwave element 7700 inaccordance with the invention. In operation, one of more digital signals7701 and, optionally, an analog signal 7702 may enter a noise isolationcircuit 7703 in a shielded structure 7704. The output of the noiseisolation circuit 7703 may drive a shielded D/A converter 7705 and theresulting analog signal from the D/A converter may tune a microwavecircuit 7706. The shielded D/A converter, in addition to a metal cover,may include a metal barrier between the D/A elements, such as resistorsor control plates to be discussed below, and the substrate to whichthese elements are attached. The tuning signal of the microwave circuitis shown as a wire 7707 in this embodiment. However, in some embodimentsof this invention, this signal is not a voltage in a wire, but a directphysical phenomenon in the tunable circuit 7706, such as a deflection ofa capacitor plate that changes the tuning. Now, several embodiments of adirect digitally controlled element in accordance with the inventionthat may be used to tune a microwave element will be described.

FIG. 8 illustrates an embodiment of a direct digitally tuned element 798that may be a digital varactor 800 in the embodiment. This varactor 800is a two-plate air-gap capacitor that may include a first cantileveredplate 801 that may be a wide cantilever beam supported on andcantilevered from an insulating foundation 802. The insulatingfoundation may be attached to a substrate 803 made of a flat rigidmaterial, such as a silicon wafer. A second plate 810 may be dividedinto a plurality of sub-plates as described below with varyingdimensions and areas that may be mounted on the substrate so that thereis an air gap between these sub-plates and the cantilevered plate. Inaccordance with the invention, the sub-plates may have differentdimensions that have the effect of adjusting the capacitance and hencethe frequency of the attached microwave device as the sub-plates arecombined together. In some embodiments due to the physical layout of themoving plate relative to the sub-plates, such as the one shown in FIG. 8in which the moving plate and sub-plates are parallel to each other, theareas of the sub-plates are related to each other in some manner whichcorresponds to the desired tuning such as the resulting frequency of thedevice connected to the direct digitally tunable element. The overallvaractor capacitance or the cantilever beam deflection. In otherembodiments, the relative capacitance generated by the sub-plates have apredetermined relationship to each other which causes the desired tuningeffect. In either case, it is the relative effect that each sub-platehas on the capacitance of the direct digitally tunable element thatcauses the desired tuning effect.

In this embodiment, a largest sub-plate 804 has dimensions and an areasuch that the sub-plate covers about ½ of the first plate 801 area(i.e., the largest sub-plate is ½ the width of the first plate is thisembodiment). The sub-plate may be positioned parallel to the cantileverbeam direction. In addition, each other sub-plate may be positionedparallel to the moving plate so that the relative effect of eachsub-plate on the capacitance of the direct digitally tunable element isrelated to the area of the sub-plate relative to the other sub-plates. Anext largest sub-plate 805 may be ½ the area of the first sub-plate 804and may be located parallel to the sub-plate 804. In addition, there maybe other sub-plates are mounted in parallel, each ½ the width of theprevious sub-plate. Thus, each of the sub-plates of the array of theseparallel sub-plates has an overlapping area with the main plate thatdiminishes by a power of 2 from the largest plate to the next plate andso on so that the overlapping areas represent binary weights. Thesmallest sub-plate may require a width below the minimum design rule ofthe particular manufacturing process so that the smallest sub-platecannot be produced. However, this limitation is overcome by reducing thelength of a smaller sub-plates 807 as shown in the figure to produce thedesired area without violating the minimum design rules.

The term “weight” as used herein refers to the magnitude of an effect.For example, a weight may describe a plate deflection in microns when aparticular sub-plate is charged, or a weight may represent the resultingcapacitance or other desired measurable effect, such as tuningfrequency, of the varactor that is applied to a tuning circuit. As longas the overall weights cause small deflections of the first plate 801,the deflection is approximately linear and the superposition of weightsis a good approximation of the combined effect. The term “binary weight”as used herein refers to the magnitude of an effect where the differencebetween the effect of different sub-plates, for example, is related by apower of 2.

Each sub-plate 804, 805, 807 may be DC-isolated from the othersub-plates and also from the substrate 803. Each sub-plate may also becharged by a binary-level (digital) voltage via one or more conductors806 connected to each sub-plate. When a selected subgroup of thesub-plates are electrically charged by the digital voltages so that apredetermined area of the sub-plates are charged, the first plate 801bends towards the sub-plates by the electrostatic attraction between thefirst plate and the sub-plates. The amount of deflection of the firstplate 801 depends how many of the sub-plates are charged (or how muchtotal area of the sub-plates is charged) so that the digital signalscontrolling the sub-plates affect the deflection of the first platewhich in turn affects the capacitance of the varactor and may be used totune a microwave device. In other words, the deflection of the firstplate changes the capacitance to ground of the variator at microwavefrequencies so that the entire collection of sub-plates 804, 805, 807acts like a ground plane at microwave frequencies providing the varactortuning capability.

While the charging of a group of the parallel sub-plate arrangementshown causes a deflection of the first plate 801 that is proportional tothe sub-plate area, this is only an approximation. Several effects maycause the proportion between the charged area of the sub-plates and therelative deflection of the first plate to be inaccurate. These effectsinclude electric field distribution and the fringe effects of capacitorplates. By proper design, the sub-plates may be adjusted in size orposition to compensate for these effects. In addition, the sub-platesmay have weights whose relative proportions are not a power of 2 and adigital circuit may compensate for the unevenness by translating thedesired total weight to the nearest combination of actual weights. Forexample, if the actual weight of sub-plates 1 to N is W₁, W₂ . . .W_(N), then it is possible to calculate the binary word B₁, B₂ . . .B_(N), where the total weight W is equal to B₁W₁+B₂W₂ . . .+B_(N)W_(N)such that this weight is the nearest to the desired weight. Each bitB_(i) where i ranges from 1 to N has binary values of 0 and VCC whereVCC is a desired controlled voltage. This concept can be directlyextended to multi-level discrete values so that B_(i) may be amulti-level valued word assuming the voltages VCC, VCC/2, VCC/4 and soon including 0, or any other set of discrete voltages. For weightsumming of W, B_(i) is assumed to have a set of discrete values 1, ½, ¼and so on including 0. The binary control word calculation may becarried out in the digital Processor 301 (shown in FIG. 3), based onexperimental or simulated estimates of the weights W₁, W₂ . . . W_(N).

As the differential deflection caused by a group of selectively chargedsubset of the sub-plates causes the first cantilevered plate 801 to flexlongitudinally, the plate 801 may be stiffened in the longitudinaldirection by placing one or more thickening bars 809 in one or moreselected places along the length of the plate 801. These bars can bemade of either insulating material or may be thick pieces of the samemetal as the plate 801. For illustration, only a single bar 809 isshown, although multiple parallel bars spaced roughly evenly along thelength of the plate 801 are preferred. Now, another embodiment of adirect digitally controlled element will be described.

FIG. 9 illustrates another embodiment of a cantilever digital varactor800. In this embodiment, the one or more sub-plates 903, 904 may beformed on or mounted on a substrate 901 perpendicular to a suspended,cantilevered plate 902 and the cantilevered plate is supported by aninsulating foundation 912. As described above, in this embodiment, therelative effect of each sub-plate on the capacitance of the capacitorand the tuning effect of the capacitor is not directly related to thearea of each sub-plate since the sub-plates closer to the end of themoving plate have a greater bending effect on the moving plate. Alargest-weight sub-plate 903 may be mounted near a free end 914 of thesuspended plate 902 and other sub-plates 904 of smaller widths may bemounted parallel to the larger sub-plate 903 and perpendicular to thecantilevered plate 902. One or more still smaller sub-plates 905 mayhave smaller overlaps with the suspended plate 902 since they areshorter to avoid violating the design rules as described above. Sincethe distance of each sub-plate 903-905 from the free end 914 of thesuspended plate 902 is not the same, the bending moments caused by thecharging of each sub-plate is different even if two sub-plates haveequal area. Therefore, for this embodiment, if binary weights aredesired, the areas or positions of the sub-plates can be adjusted bysimulation or experimentation as will now be described.

In accordance with the invention, a method for adjusting the area orposition of each sub-plate may include positioning a first sub-plate ofa desired weight-effect relative to the cantilevered plate where desiredand adjusting the width of said plate until the desired weight-effectassociated with said sub-plate is achieved and confirmed by measurementor simulation. Next, a second sub-plate may be positioned relative tothe first sub-plate and the dimensions of the second sub-plate (i.e.,length, width and/or thickness) may be adjusted until the desiredweight-effect associated with said second sub-plate is achieved andconfirmed by measurement or simulation. Returning to the varactor shownin FIG. 9 as an example, the first sub-plate 903 may be positioned wheredesired relative to the cantilever plate 902 and may be widened until ½of the total weight (i.e., ½ the total deflection of the cantileverplate is caused by the first sub-plate) is achieved. Then the secondplate 904 is positioned and width-adjusted until an extra ¼ of weight isadded. The process continues with a third sub-plate at ⅛^(th) of theweight and so on until reaching a plate too narrow to manufacture. Tomake the sub-plates that are too narrow, these sub-plates are madeshorter rather than narrower as described above. The operation of thisembodiment is the same as above and will therefore not be describedhere.

Although an array of sub-plates parallel to the cantilever plate chargedwith digital voltages (FIG. 8) and an array of sub-plates perpendicularto the cantilever plate charged with digital voltages (FIG. 9) are shownand described, other combinations of sub-plates are also possible inaccordance with the invention. For example, one of the sub-plates may becharged with an analog voltage, such as small sub-plate 906 in FIG. 9.

Returning to FIG. 9, each sub-plate can be connected by a conductor 907to a bonding pad 908 if so desired. The suspended plate 902 may be alsoconnected to a bonding pad 910 via a printed transmission line 909 to beable to apply a voltage to the plate or measure the deflection orcapacitance of the first plate. In many applications, the cantileverplate may be electrically connected directly or via bond wire to aresonating element that it is supposed to tune, such as in FIG. 13 asdescribed below.

The digitization of micro-machined capacitors by use of multiple controlelectrodes (sub-plates) in accordance with the invention is applicableto other forms of capacitors, including interdigital capacitors. Thus.similar to the parallel and perpendicular sub-plate arrangements shownin FIGS. 8 and 9, respectively, similarly divided sub-plates may be usedfor an interdigital capacitor as shown in FIGS. 10 and 11 and as willnow be described.

FIG. 10 illustrates a plate arrangement in accordance with the inventionfor an interdigital capacitor 1000. An electrically conductivecantilever beam 1001 may be integrally connected to a set of conductiveparallel plates 1002 that move as the cantilever plate deflects and thecantilevered plate is mounted on and cantilevered from an insulativesupport-base 1003. One or more static plates 1004 are attached to thesubstrate and each static plate is positioned between two conductiveparallel plates 1002. These static plates are conductive, but areelectrically isolated from each other at DC and low frequency. However,at microwave frequencies, they are at ground potential. The movement ofthe cantilevered plate 1001 is caused by electrostatic attraction, likethe cantilever example of FIG. 8. The binary weights for tuning arecreated by grouping one or more static plates to represent a desiredweight. In the example shown, four central static plates 1010 areelectrically connected by a conductor 1005 together, forming the mostsignificant bit (MSB), while another pair is connected by a conductor1006 to form the next bit. In this embodiment, the minimum number ofplates per bit is two, to allow symmetry of the attraction forces. Toadd more bits, other pairs of plates are made shorter in the horizontaldimensions, i.e. the plates connected to a wire 1007 may be shorter thanthe plates connected to the wire 1006.

If the moving plates 1002 are not rigid enough to avoid sticking to astatic plate 1004 in the boundary between two static plates of differentbits, another plate arrangement can be used, as shown in FIG. 11 and asdescribed below. While the drawing in FIG. 10 shows a cantileverstructure 1001 that deflects vertically due to electrostatic attraction,as indicated by an arrow 1008, other well known interdigital capacitorsexist in which the deflection movement is horizontal, as indicated byanother arrow 1009. Since the capacitance (and hence the tuning) issubstantially proportional to the overlapping areas of the static platesand moving plates, either a horizontal or vertical deflectioninterdigital capacitor may be used in accordance with the invention toprovide microwave tuning.

FIG. 11 shows another embodiment of a direct digitally tunedinterdigital capacitor 1100 in accordance with the invention. In thisembodiment, a set of static plates 1101, 1102 may be divided over theirlengths into one or more sub-plates. For example, a MSB sub-plate 1101may be sub-divided into on or more sub-plates such as sub-plates 11084,1109. The MSB plate 1101 may also be electrically connected to othersimilar sub-plates, such as the sub-plate 1102. All of these sub-platesmay then be electrically connected to a conductor 1103. After theminimum attainable plate width has been reached, other sub-plates 1104may be mounted on the external side only and at a varying distance froma static plate 1106, as shown with the sub-plate 1105. These sub-platesare mounted symmetrically on both sides of the moving structure, so thatsub-plate 1104 has a peer 1107.

FIG. 12 illustrates a cross-sectional view of a digitally controlledcapacitor 1200 in accordance with a preferred embodiment of theinvention. In a preferred embodiment, the digitally controlled capacitoris manufactured on a semiconductor substrate taking advantage ofexisting semiconductor manufacturing processes available in themicroelectronic industry. This capacitor 1200 uses a set of sub-platesperpendicular to the cantilever plate discussed above in conjunctionwith FIG. 9. The capacitor 1200 may include a semiconductor substrate1201, such as silicon or gallium arsenide, that supports the capacitorstructure. A metalization layer 1202 creates a shield between thecapacitor and any noise present in the substrate. A set of sub-plates1205 may be etched from a metalization layer resting on top of aninsulating layer 1206. Next, a capacitor cantilever plate 1203 may bedeposited on an insulation layer 1204. The suspension is manufactured bya well known air-bridge technique in which the air gap was initially asacrificial layer on which the cantilever plate 1203 was deposited andetched, and finally the sacrificial layer was removed. Various wellknown wafer processing techniques may allow the deposition of othercomponents found in integrated circuits, including resistors 1207 andtheir interconnection leads 1208. Various active semiconductor devices,such as a MOS transistor 1209, may also be fabricated. The choice of thetypes of active devices is based on the technology available for aparticular substrate material. The cantilever plate 1203 is notnecessarily connected to the active device 1209 as illustrated in FIG.12.

While the substrate 1201 is made of a semiconductor material in thepreferred embodiment, other substrate materials are possible, such as aceramic or a metal. If the substrate is made of metal, the shield layer1202 may consist of another type of metal to reduce noise. Anotheralternative is to have the substrate and shield made of the same metal,essentially obviating the need for the separate layer 1202.

FIG. 13 illustrates a digitally controlled cantilever plate capacitor1301 in accordance with the invention that may be electrically connectedto a passive resonator element 1302. The resonator element 1302 may bemade of a conducting surface that may be a microstrip trace on asubstrate or a metal structure suspended in air and bonded to thesubstrate by hybrid technologies. The capacitor 1301 may be connected toresonator 1302 by a bond wire 1303 or regular printed conductor thinfilm techniques. Now, another embodiment of a direct digitallycontrolled microwave element will be described.

FIG. 14 illustrates another embodiment for directly digitallycontrolling a microwave tuning element that may be part of a microwavecircuit 1409, such as oscillator and filter. The digitally controlledmicrowave circuit may be enclosed in a shielded structure 1401. Theshield 1401 may contain, for example, the DCO 603 shown in FIG. 6 or theDCO 704 shown in FIG. 7. This shield 1401 should not be confused withthe shield 604 of the outer control structure. The clean logic signals,such as the MSB 1402 and analog voltage or LSB 1403, are fed to acontrolled device 1404, such as a variable capacitor, via an R-2R D/Aresistor ladder converter.

The resistors shown in FIG. 14 may all be of the same value. The 2Rbranch for each bit, such as MSB 1402, may split equally between aresistor 1405 outside the shield and a resistor 1406 inside the shield,and between them a noise rejection bypass capacitor 1407, implementedpartially or entirely as a feed-through capacitor. Each resistor pair1405, 1406 may consist of unequal resistors as long as their sum equalsthe 2R value of the DC-equivalent resistor ladder structure. It ispossible even to have the entire branch resistance 2R kept outside ofthe shield for maximum RC time constant formed by the resistor 1405 andcapacitor 1407. In that case, the resistor 1406 can be replaced by aconductor. The DC control voltage is isolated at the microwave frequencyby an RFC 1408, and the control voltage drives the voltage tuned device,such as a varactor 1404. This varactor may consist of a conventionalsemiconductor tuning diode or a micro-machined varactor. The tuningdevice 1404 tunes the microwave circuit 1409, such as an oscillator or afilter structure.

FIG. 15 illustrates a three dimensional cut-away view of a shield 1501.In particular, the shield 1501 may consist of a metal cap 1509 bonded toa metal layer 1502 on top of a substrate 1503. One or more resistors1504 may be formed on an insulation layer 1505 above the metal layer1502. One or more bonding pads 1506 may provide connection to theoutside components. Thus, in this embodiment, the shield 1401 of FIG. 14is implemented as a combination of the cap 1509 and the metal layer1502. The insulation layer 1505 may include conductive via holes 1507.The components shown in FIG. 15 may correspond to various components inFIG. 14. For example, the resistor 1504 may correspond to the resistor1406, while the resistor 1508 may represent the combined resistance ofthe two resistors 1411. The feed-through capacitor 1407 may be thecapacitance between the input conductor 1509 and the ground structuressurrounding it, consisting of the shield 1501 and the conductive layer1502 underneath.

The manufacturing of the various structures disclosed above is possibleby several existing techniques related to thin film technology andmicromachining. Some references to literature on microwave tunablecapacitors can be found in the book “Introduction toMicroelectromechanical (MEM) Microwave Systems” by Hector De Los Santos,Artech House, 1999, and “Fundamentals of Microfabrication” by MarcMadou, CRC Press, 1997.

FIG. 16 illustrates a typical radio transceiver incorporating a directdigitally controlled devices techniques in accordance with the inventiondisclosed above that improve a radio transceiver front end. This figureshows various elements the correspond to elements in FIG. 3 and theseelements will not be described here, but the digitally controlledelements will be pointed out. A digital processor 301 may tune thefilters 307, 311 digitally via control buses 1604. Each bus, drawn as athick line, may also contain an analog line, in which case, the digitalprocessor includes a D/A converter 317 either built in or mountedexternally. The diplexer 308 may be controlled by a transmit bus 1606and a receive bus 1607. The DCO 305 refers to the “outer structure”,such as the entire circuitry shown in FIG. 7, in which the control bit705 corresponds to a wire in a VCO control bus 1609. The coarse tuningof any of these filers and oscillators may be done by a known mappingbetween the frequency and the tuning control word. Such a mapping can befound by design or product calibration and later fine tuning can be doneby feedback techniques, such as the phase locked loop for the DCO 305and receiver signal level optimization for the diplexer 308 and receivefilter 311. To provide feedback control of the transmit filter 307 anddiplexer 308, an amplitude measurement feedback line 1611 may beincluded. Feedback control is accomplished by searching for the optimumdesired level. If desired, dither techniques of turning on and off oneof the lowest significance control bits can be used to search for thecontrol loop direction to the maximum value.

While the foregoing has been with reference to a particular embodimentof the invention, it will be appreciated by those skilled in the artthat changes in this embodiment may be made without departing from theprinciples and spirit of the invention, the scope of which is defined bythe appended claims.

1. A digital microwave transceiver, comprising: means for receiving asignal to be transmitted using a microwave frequency; and means forupconverting the signal onto a predetermined microwave frequency, theupconverting subsystem includes one or more direct digitally tunedcircuits for tuning the transceiver wherein each digitally timed circuitcomprises a microwave circuit that is being tuned to a predeterminedmicrowave frequency, an electrically tunable element for tuning themicrowave circuit to the predetermined microwave frequency, theelectrically tunable element being attached to a substrate and furthercomprising a plurality of tuning signals, means for noise filtering saidtuning signals to generate noise filtered tuning signals, a converter,based on the noise filtered tuning signals, that controls the frequencyof the microwave circuit to a predetermined microwave frequency and ananalog tuning signal that is integrated into the substrate and mountedon said noise filtering means.
 2. The transceiver of claim 1, whereinthe converter further comprises a digital to analog converter thatchanges a predetermined characteristic in response to the noise filtereddigital signals so that the microwave circuit tunes itself to amicrowave frequency based on the digital signals.
 3. The transceiver ofclaim 2, wherein said digital to analog converter further comprises oneor more resistors in a resistor ladder wherein at least some of theresistors are located inside of a noise shield and some of the resistorsare located outside of the noise shield, wherein the resistors insideand outside of the noise shield are connected together by a noiserejection bypass capacitor that is integrated into the noise shield. 4.The transceiver of claim 1, wherein said tuning signals comprise one ormore digital signals for effecting the coarsely tuning of the microwavecircuit and at least one analog tuning signal for effecting the finetuning of the microwave circuit.
 5. The transceiver of claim 1, whereinsaid converter comprises a capacitor including at least one movingcapacitor plate and two or more sub-plates electrically isolated at DCor low frequencies from each other and from the moving plate forelectrically controlling the deflection of the moving plate based on thetuning signals.
 6. The transceiver of claim 5, wherein the sub-plates ofthe capacitor further comprises two or more sub-plates, a firstsub-plate having an area such that the capacitance of the capacitor isapproximately ½ of the total capacitance of the capacitor and a secondsub-plate having an area equal to approximately ½ of the area of firstsub-plate so that when one or more of the sub-plates are charged, thecontrol plate deflects a predetermined amount to change the capacitanceof the capacitor and tone the microwave device attached to thecapacitor.
 7. The transceiver of claim 1, wherein the tuning signals areconnected to the sub-plates of the capacitor and wherein the sub-platesof the capacitor further comprises two or more sub-plates, a firstsub-plate having predetermined area such that the capacitance change ofthe capacitor based on the first sub-plate is approximately ½ of thetotal capacitance of the capacitor and a second sub-plate having asecond predetermined area so that the change in the capacitance of thecapacitor based on the second sub-plate is approximately ½ the changecaused by the first sub-plate to create a binary weighting of thesub-plates so that when one or more of the sub-plates are changed, thecontrol plate deflects a predetermined amount to change the capacitanceof the capacitor and tune the microwave device attached to thecapacitor.